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MIC3230/1/2 constant current boost controller for driving high power leds bringing the power to light? bringing the power to light is a trademark of micrel, inc. microlead frame and mlf are registered trademark of amkor technologies. micrel inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? te l +1 (408) 944-0800 ? fax + 1 (408) 474-1000 ? http://www.micre l.com general description the MIC3230/1/2 are constant current boost switching controllers specifically designed to power one or more strings of high power leds. the MIC3230/1/2 have an input voltage range from 6v to 45v and are ideal for a variety of solid state lighting applications. the MIC3230/1/2 utilizes an ex ternal power device which offers a cost conscious solution for high power led applications. the powerful drive circuitry can deliver up to 70w to the led system. power consumption has been minimized through the implementation of a 250mv feedback voltage reference providing an accuracy of 3%. the mic323x family is dimmable via a pulse width modulated (pwm) input signal and also features an enable pin for low power shutdown. multiple MIC3230 ics can be synchronized to a common operating frequency. the cloc ks of these synchronized devices can be used together in order to help reduce noise and errors in a system. an external resistor sets the adjustable switching frequency of the MIC3230/1. the switching frequency can be between 100khz and 1mhz. setting the switching frequency provides the mechanism by which a design can be optimized for efficiency (performance) and size of the external components (cost). the mic323x family of led drivers also offer the following protection features: over voltage protection (ovp), thermal shutdown and under- voltage lock-out (uvlo). the mic3231 offers a dither feature to assist in the reduction of emi. this is particularly useful in sensitive emi applications, and provides for a reduction or emissions by approximately 10db. the mic3232 is a 400khz fixed frequency device offered in a small msop-10 package. the MIC3230/1 are offered in both the epad tssop-16 package and the 3mm 3mm mlf ? -12 package. data sheets and support documentation can be found on micrel?s web site at: www.micrel.com. features ? 6v to 45v input supply range ? capable of driving up to 70w ? ultra low emi via dithering on the mic3231 ? programmable led drive current ? feedback voltage = 250mv 3% ? programmable switching frequency (MIC3230/1) or 400khz fixed frequency operation (mic3232) ? pwm dimming and separate enable shutdown ? frequency synchronization with other MIC3230s ? protection features: over voltage protection (ovp) over temperature protection under-voltage lock-out (uvlo) ? packages: iadj is6 5 1vin en pwmd comp 10 vdd drv pgnd ovp 9 8 7 2 3 4 iadj is vin en pwmd comp 1 vdd drv pgnd ovp 2 3 4 5 fs epad sync/nc 6 8 12 11 10 9 7 1n/c vin en pwmd comp iadj fs agnd 16 n/c vdd drv pgnd ovp is sync/nc n/c 15 14 13 12 11 10 9 2 3 4 5 6 7 8 epad mic3232 msop-10 MIC3230/1 mlf-12 MIC3230/1 tssop-16 ? ?40c to +125c junction temperature range applications ? street lighting ? solid state lighting ? general illumination ? architectural lighting ? constant current power supplies _________________________________________________________________________________________________ january 2009 m9999-011409-a (408) 955-1690
micrel, inc. MIC3230/1/2 january 2009 2 m9999-011409-a typical application l 47h d1 r8 100k r9 4.33k cout 4.7f 100v r2 100k radj 1/4w rfs 16.5k ccomp 10nf cin 4.7f/50v rslc 51 v fb = 0.25v rcs 1/2w analog ground power ground vout vin pwmd enable synch to other MIC3230 iled return led 1 led x q1 comp pwmd vdd agnd pgnd epad is ovp drv vin en iadj MIC3230/31 sync fs c3 10f 10v figure 1. typical application of the MIC3230 led driver product option matrix MIC3230 mic3231 mic3232 input voltage 6v to 45v 6v to 45v 6v to 45v synchronization yes no no dither no yes no frequency range adj from 100khz to 1mhz adj from 100khz to 1mhz fixed freq. = 400khz package epad tssop-16 3mm 3mm mlf ? -12 epad tssop-16 3mm 3mm mlf ? -12 msop-10 ordering information part number temperature range package lead finish MIC3230ytse ?40 to +125c epad tssop-16 pb-free MIC3230yml ?40 to +125c 3mm x 3mm mlf ? -12l pb-free mic3231ytse ?40 to +125c epad tssop-16 pb-free mic3231yml ?40 to +125c 3mm x 3mm mlf ? -12l pb-free mic3232ymm ?40 to +125c msop-10 pb-free
micrel, inc. MIC3230/1/2 january 2009 3 m9999-011409-a pin configuration iadj is6 5 1vin en pwmd comp 10 vdd drv pgn d ovp 9 8 7 2 3 4 iadj is vin en pwmd comp 1 vdd drv pgnd ovp 2 3 4 5 fs epad sync/nc 6 8 12 11 10 9 7 1n/c vin en pwmd comp iadj fs agnd 16 n/c vdd drv pgnd ovp is sync/nc n/c 15 14 13 12 11 10 9 2 3 4 5 6 7 8 epad msop-10 (mm) mic3232 3mmx3mmmlf ? -12l (ml) MIC3230, mic3231 see product option matrix for selection tssop-16 (tse) MIC3230, mic3231 see product option matrix for selection pin description pin number 3x3mlf pin number tssop-16l pin number msop-10l pin name pin function -- 1 -- nc no connect 1 2 1 vin input voltage (power) 6v to 45v 2 3 2 en enable control (input). logic high ( 1.5v) enables the regulator. logic low ( 0.4v) shuts down the regulator. connect a 100k ? resistor from en to vin. 3 4 3 pwmd pwm input. high signal terminates the output power. low signal starts up the output power. 4 5 4 comp compensation (output): for external compensation 5 6 5 iadj feedback (input) 6 7 -- fs frequency select (input). connected to a resistor to determine the operating frequency -- 8 -- agnd analog ground -- 9 -- nc no connect 7 10 -- sync sync (output). connect to another MIC3230 to synchronize multiple converters. 8 11 6 is current sense (input). connected to external current sense resistor which in turn is connected to the source of the external fet as well as an external slope compensation resistor 9 12 7 ovp ovp divider connection (output). connect the top of the divider string to t he output. if the load is disconnected, the output voltage will rise until ovp reaches 1.25v and then will regulate around this point 10 13 8 pgnd power ground 11 14 9 drv drive output: connect to the gate of external fet (output) 12 15 10 vdd vdd filter for internal power rail. do not connect an external load to this pi n. connect 10f to gnd. -- 16 -- nc no connect -- -- -- epad connect to agnd
micrel, inc. MIC3230/1/2 january 2009 4 m9999-011409-a absolute maximum ratings (1) supply voltage (v in ) .....................................................+48v enable pin voltage ........................................... -0.3v to +6v iadj voltage ..................................................................+6v lead temperature (solde ring, #sec .) ......................... 260c storage temperature (ts) ..........................-65c to +150c esd rating (3) ..................... MIC3230= 1500v hb, 100vmm .........................................mic3232= 2kv hb, 100vmm .................................... mic3231= 1500v hb, 150vmm operating ratings (2) supply voltage (v in )......................................... +6v to +45v junction temperature (t j)........................ ?40c to +125c junction thermal resistance msop-10 ( ja ) ..............................................130.5c/w epad tssop-16 ( ja ) ...................................36.5c/w 3mmx3mm mlf ? -12l ( ja ).............................60.7c/w electrical characteristics (4) v in = 12v; v en = 3.6v; l = 47h; c = 4.7f; t j = 25c, bold values indicate ?40c ? t j ? +125c, unless noted. symbol parameter condition min typ max units v in supply voltage range 6 45 v uvlo under voltage lockout 3.5 4.9 5.5 v i vin quiescent current v fb > 275mv (to ensure device is not switching) 3.2 10 ma i sd shutdown current v en = 0v 30 a room temperature (3%) 242.5 250 257.5 mv v iadj feedback voltage (at iadj) ?40c ? t j ? +125c (5%) 237.5 250 262.5 mv i adj feedback input current v fb = 250mv 1.2 3 a line regulation v in = 12v to 24v 2 % load regulation v out to 2 v out 2 % d max maximum duty cycl e MIC3230 & mic3232 mic3231 90 88 % % v en enable threshold turn on turn off 1.5 1.15 1.1 0.4 v v i en enable pin current ven = 3.3v ren = 100k 17 30 a v pwm pwmd threshold turn on turn off 1.5 0.75 0.7 0.4 v v f pwmd pwmd frequency range note 5 (l = 47h; c = 4.7f) 0 500 hz f sw programmable oscillator frequency r freq = 82.5k r freq = 21k r freq = 8.25k 360 109 400 950 440 khz khz khz f sw fixed frequency option (mic3232ymm) 360 400 440 khz f dither low emi (mic3231) frequency dither shift from nominal 12 % v sens current limit threshold voltage r sense = 390 0.315 0.45 0.585 v i sense i sense peak current out r sense = 390 250 a v ovp over voltage protection 1.203 1.24 1.277 v driver impedance sink source 2.4 2 3.5
micrel, inc. MIC3230/1/2 january 2009 5 m9999-011409-a v drh driver voltage high v in = 12v 7 9 11 v t j over-temperature threshold shutdown 150 c thermal shutdown hysteresis 5 c notes: 1. exceeding the absolute maximum rating may damage the device. 2. the device is not guaranteed to function outside its operating rating. 3. devices are esd sensitive. handling precautions recommended. human body model, 1.5k ? in series with 100pf. 4. specification for packaged product only. 5. guaranteed by design
micrel, inc. MIC3230/1/2 january 2009 6 m9999-011409-a typical characteristics
micrel, inc. MIC3230/1/2 january 2009 7 m9999-011409-a 11.8 11.85 11.9 11.95 12 12.05 12.1 12.15 12.2 0 25 50 75 100 125 150 output voltage (v) load (ma) load regulation v in = 3.6v
micrel, inc. MIC3230/1/2 january 2009 8 m9999-011409-a functional description a constant output current converter is the preferred method for driving leds. sma ll variations in current have a minimal effect on the light output, whereas small variations in voltage have a significant impact on light output. the mic323x family of led drivers are specifically designed to operate as constant current led drivers and the typical application schematic is shown in figure 1. the mic323x family are designed to operate as a boost controller, where the output voltage is greater than the input voltage. this configuration allows for the design of multiple leds in series to help maintain color and brightness. the mic323x family can also be configured as a sepic controller, where the output voltage can be either above or below the input voltage. the MIC3230/1/2 have a very wide input voltage range, between 6v and 45v, to help accommodate for a diverse range of input voltage applications. in addition, the led current can be programmed to a wide range of values through the use of an external resistor. this provides design flexibility in adjusting the current for a particular application need. the MIC3230/1/2 features a low impedance gate driver capable of switching large mosfets. this low impedance helps provide higher operating efficiency. the mic323x family can control the brightness of the leds via its pwm dimming c apability. applying a pwm signal (up to 500hz) to the pwmd pin allows for control of the brightness of the led. each member of the mic323x family employs peak current mode control. peak current mode control offers advantages over voltage mode control in the following manner. current mode control can achieve a superior line transient performance compared to voltage mode control and through small signal analysis (not shown here), current mode control is easier to compensate than voltage mode control, thus allowing for a less complex control loop stability design. figure 2 shows the functional block diagram. control rslc rcs vdd ovp pwmd gnd q1 c5 l1 is s r q clock out vramp out sync sync fs iadj radj en ldo vin internal vbias vdd fs v c v s 10k 0.25v ai ai = 1.4v vramp in islope comp out islope comp leading edge blanking 0.45v t t 250a/t vout comp vin led led drv vout figure 2. MIC3230 functional block diagram
micrel, inc. MIC3230/1/2 january 2009 9 m9999-011409-a power topology constant output current controller the mic323x family are peak current mode boost controllers designed to drive high power leds. unlike a standard constant output volt age controller, the mic323x family has been designed to provide a constant output current. the mic323x family is designed for a wide input voltage range, from 6v to 45v. in the boost configuration, the output can be set from v in up to 100v. as a peak current mode controller, the mic323x family provides the benefits of superior line transient response as well as an easier to design compensation. this family of led drivers features a built-in soft-start circuitry in order to prevent start-up surges. other protection features include: ? current limit (i limit ) - current sensing for over current and overload protection ? over voltage protection (ovp) - output over voltage protection to prevent operation above a safe upper limit ? under voltage lockout (uvlo) ? uvlo designed to prevent operation at very low input voltages setting the led current the current through the led string is set via the value chosen for the current sense resistor, r adj . this value can be calculated using equation 1: eq. (1) adj led r v i 25.0 = another important parameter to be aware of in the boost controller design, is the ripple current. the amount of ripple current through the led string is equal to the output ripple voltage divided by the led ac resistance (r led ? provided by the led manufact urer) plus the current sense resistor (r adj ). the amount of allowable ripple through the led string is dependent upon the application and is left to the designer?s discretion. this equation is shown in equation 2: eq. (2) ) ( adj led out led rr v i ripple + ? where out led out c tdi v ripple = reference voltage the voltage feedback loop of the mic323x uses an internal reference voltage of 0.25v with an accuracy of 3%. the feedback voltage is the voltage drop across the current setting resistor (r adj ) as shown in figure 1. when in regulation the voltage at i adj will equal 0.25v. output over voltage protection (ovp) the mic323x provides an ovp circuitry in order to help protect the system from an overvoltage fault condition. this ovp point can be programmed through the use of external resistors (r8 and r9 in figure 1). a reference value of 1.245v is used for the ovp. equation 3 can be used to calculate the resistor value for r9 to set the ovp point. eq. (3) 1)245.1/( 8 9 ? = ovp v r r led dimming the mic323x family of led drivers can control the brightness of the led string via the use of pulse width modulated (pwm) dimming. a pwm input signal of up to 500hz can be applied to the pwm dim pin (see figure 1) to pulse the led string on and off. it is recommended to use pwm dimming signals above 120hz to avoid any recognizable flicker by the human eye. pwm dimming is the preferred way to dim a led in order to prevent color/wavelength shifting, as occurs with analog dimming. the output current level remains constant during each pwmd pulse. oscillator and switching frequency selection the mic323x family features an internal oscillator that synchronizes all of the switching circuits internal to the ic. this frequency is adjustable on the MIC3230 and mic3231 and fixed at 400khz in the mic3232. in the MIC3230/1, the switching frequency can be set by choosing the appropriate value for the resistor, r 1 according to equation 4: eq. (4) 035.1 )( 7526 )( ? ? ? ? ? ? ? ? = khzf kr sw fs sync (MIC3230 only) multiple MIC3230 ics can be synchronized by connecting their sync pins together. when synchronized, the MIC3230 with the highest frequency (master) will override the other MIC3230s (slaves). the internal oscillator of the master ic will override the os cillator of the slave part(s) and all MIC3230 will be synchronized to the same master switching frequency. the sync pin is designed to be used only by other MIC3230s and is available on the MIC3230 only. if the sync pin is being unused, it is to be left floating (open). in the mic3231, the sync pin is to be left floating (open).
micrel, inc. MIC3230/1/2 january 2009 10 m9999-011409-a dithering (mic3231 only) the mic3231 has a feature wh ich dithers the switching frequency by 12%. the purpose of this dithering is to help achieve a spread spectr um of the conducted emi noise. this can allow for an overall reduction in noise emission by approximately 10db. internal gate driver external fets are driven by the mic323x?s internal low impedance gate drivers. these drivers are biased from the v dd and have a source resistance of 2 ? and a sink resistance of 3.5 ? . v dd v dd is an internal linear regulator powered by v in and v dd is the bias supply for the inter nal circuitry of the mic323x. a 10f ceramic bypass capacitor is required at the v dd pin for proper operation. this pin is for filtering only and should not be utilized for operation. current limit the mic323x family features a current limit protection feature to prevent any current runaway conditions. the current limit circuitry monitors current on a pulse by pulse basis. it limits the current through the inductor by sensing the voltage across r cs . when 0.45v is present at the is pin, the pulse is truncated. the next pulse continues as normally until the is pin reaches 0.45v and it is truncated once again. this will continue until the output load is decreased. select r cs using equation 5: eq. (5) limit pk min max l sw in out cs i fl dvv r _ 45.0 u u slope compensation the mic323x is a peak current mode controller and requires slope compensation. slope compensation is required to maintain internal stability across all duty cycles and prevent any unstable oscillations. the mic323x uses slope compensation that is set by an external resistor, r slc . the ability to set the proper slope compensation through the use of a single external component results in design flexibility. this slope compensation resistor, r slc , can be calculated using equation 6: eq. (6) sw cs in out slc fal rv v r min max uu u p 250 where v in_max and v out_max can be selected to system specifications. current sense i s the is pin monitors the rising slope of the inductor current (m1 in figure 5) and also sources a ramp current (250a/t) that flows through r slc that is used for slope compensation. this ramp of 250a per period, t, generates a ramped voltage across r slc and is labeled v a in figure 3. the signal at the is pin is the sum of v cs + v a (as shown in figure 3). the current sense circuitry and block diagram is displayed in figure 4. the is pin is also used as the current limit (see the previous section on current limit). figure 3. slope compensation waveforms soft start the boost switching convertor features a soft start in order to power up in a controlled manner, thereby limiting the inrush current from the line supply. without this soft start, the inrush current could be too high for the supply. to prevent this, a soft start delay can be set using the compensation capacitor (c comp in figure 1). for switching to begin, the voltage on the compensation cap must reach about 0.7v. switching starts with the minimum duty cycle and increases to the final dut y cycle. as the duty cycle increases, v out will increase from v in to it?s final value. a 6a current source charges the compensation capacitor and the soft start time can be calculated in equation 7: eq. (7) 6 u | v comp_steady_state is usually between 0.7v to 3v, but can be as high as 5v. eq. (8) pk a state steady comp vcsvai v pk u _ _ where: tdr t i v slc ramp a pk uuu and cs pk lcs riv pk u _ ai = 1.4 v/v d = duty cycle (0 to1) t = period a 10nf ceramic capacitor will make this system stable at all operating conditions.
micrel, inc. MIC3230/1/2 january 2009 11 m9999-011409-a eq. (10) () () 12 2 _ 2 _ _ ppin rmsin avein i i i ? = leading edge blanking large transient spikes due to the reverse recovery of the diode may be present at the leading edge of the current sense signal. (note: drive cu rrent can also cause such spikes) for this reason a switch is employed to blank the first 100ns of the current sense signal. see figure 6. eq. (11) 2 _ _ _ ppin avein peakin i i i + = note: if i in_pp is small then i in_ave nearly equals i in_rms eq. (9) in out out rmsin veff iv i = _ v a +rslc? is s r q 0.45v 0.45v c comp comp r comp = 10k v c pwm comparator iadj v a = i ramp r slp current limit clock 250a/t vin drv l1 d1 v cs il v cs = il r cs r cs + ? ai figure 4. current sense circuit (an explanation of the is pin) cloc k pwm v c v c v c i fet i diode 0 0 0 i l i l_ave = i in_ave i l_pk = i l_ave + 1/2 i l_pp i out t dt (1-d)t i l_pp i fet_rms i l_ave = i in_ave m1 m2 figure 5. current waveforms
micrel, inc. MIC3230/1/2 january 2009 12 m9999-011409-a figure 6. is pin and v rcs (ch1 = switch node, ch2 = is pin, ref1 = v cs ) design procedure for a led driver symbol parameter min nom max units input v in input voltage 8 12 14 v i in input current 2 a output leds number of leds 5 6 7 v f forward voltage of led 3.2 3.5 4.0 v v out output voltage 16 21 28 v i led led current 0.33 0.35 0.37 a i pp required i ripple 40 ma pwmd pwm dimming 0 100 % ovp output over voltage protection 30 v system f sw switching frequency 500khz eff efficiency 80 % v diode forward drop of schottky diode 0.6 v table 2. design example parameters
micrel, inc. MIC3230/1/2 january 2009 13 m9999-011409-a l 47h d1 r8 100k r9 4.33k cout 4.7f 100v r2 100k radj 1/4w rfs 16.5k ccomp 10nf cin 4.7f/50v rslc 51 v fb = 0.25v rcs 1/2w analog ground power ground vout vin pwmd enable synch to other MIC3230 iled return led 1 led x q1 comp pwmd vdd agnd pgnd epad is ovp drv vin en iadj MIC3230/31 sync fs c3 10f 10v figure 7. design example schematic design example in this example, we will be designing a boost led driver operating off a 12v input. this design has been created to drive six leds at 350ma with a ripple of about 12%. we are designing for 80% efficiency at a switching frequency of 500khz. select r fs to operate at a switching frequency of 500khz, the r fs resistor must be chosen using equation 3. () () = = k kr fs 6.16 500 7526 035.1 use the closest standard value resistor of 16.5k ? . select r adj having chosen the led drive current to be 350ma in this example, the current can be set by choosing the radj resistor from equation 1: == 71.0 35.0 25.0 a v r adj the power dissipation in this resistor is: () mw rirp adj adj 87 * 2 = = use a resistor rated at ? watt or higher. choose the closest value from a resistor manufacture. operating duty cycle the operating duty cycle can be calculated using equation 12 provided below: eq. (12) diode diode vvout vvineffvout d + +? = ) ( these can be calculated for the nominal (typical) operating conditions, but should also be understood for the minimum and maximum system conditions as listed below. schottky nom schottky nom nom nom v vout vvineff vout d + +? = ) ( schottky schottky v vout vvineff vout d + +? = max min max max ) ( schottky schottky v vout vvineff vout d + +? = min max min min ) ( therefore d nom =56% d max = 78% and d min = 33% inductor selection first, it is necessary to calculate the rms input current (nominal, min and max) for the system given the operating conditions listed in the design example table. this minimum value of the rms input current is necessary to ensure proper operation. using equation 9, the following values have been calculated: rmsa veff i v i in out out rmsin _64.1 min_ max_ max_ max__ = =
micrel, inc. MIC3230/1/2 january 2009 14 m9999-011409-a rmsa veff i v i nomin nom out nom out nom rmsin _78.0 _ _ _ __ = = rmsa veff i v i in out out rmsin _48.0 max_ min_ min_ min__ = = iout is the same as iled selecting the inductor current (peak-to-peak), i l_pp , to be between 20% to 50% of i in_rms_nom , in this case 40%, we obtain: pp nomrmsin nomppin a i i ? = = = 31.078.0*4.0 4.0 __ __ (see the current waveforms in figure 5). it can be difficult to find large inductor values with high saturation currents in a surface mount package. due to this, the percentage of the ripple current may be limited by the available inductor. it is recommended to operate in the continuous conduction mode. the selection of l described here is for continuous conduction mode. eq. (13) ppin in i tdv l _ = using the nominal values, we get: h a s v l 43 31.0 256.012 = = select the next higher standard inductor value of 47h. going back and calculating the actual ripple current gives: eq. (13a) pp nom nomin ppin a uh us v l td v i 29.0 47 256.012 _ _ = = = the average input current is different than the rms input current because of the ripple current. if the ripple current is low, then the average input current nearly equals the rms input current. in the case where the average input current is different than the rms, equation 10 shows the following: eq. (13b) () ( ) 12 2 _ 2 max__ max__ ppin rmsin avein i i i ? = ()() a i avein 64.112/29.064.1 2 2 max__ ? = the maximum peak input current i l_pk can found using equation 11: a i i i ppl avein pkl 78.1 5.0 max__ max__ max__ = + = the saturation current (i sat ) at the highest operating temperature of the inductor must be rated higher than this. the power dissipated in the inductor is: eq. (13c) dcr i p rmsin inductor = 2 max__ current limit and slope compensation having calculated the i l_pk above, we can set the current limit 20% above this maximum value: aa i limit pkl 9.16.12.1 _ == the internal current limit comparator reference is set at 0.45v, therefore when , the ic enters current limit. 45.0 _ = pinis v eq. (14) ( ) pk a vcsv pk + = 45.0 where is the peak of the waveform and is the peak of the vcs waveform pk a v a v pk vcs eq. (14a) cs pkl slc ramp ridri limit + = _ 45.0 to calculate the value of the slope compensation resistance, r slc , we can use equation 5: ( ) sw cs in out slc fal rv v r min max ? = 250 first we must calculate r cs , which is given below in equation 15: eq. (15) () limit pkl sw min max cs i fl d vin vout r _ max 45.0 + ? = therefore; ()() = + ? = m a khz h vv r cs 179 9.1 500 47 50.0828 45.0 using a standard value 150m ? resistor for r cs , we obtain the following for r slc : ( ) = ?? = 511 500 250 47 150828 khzah m r slc ? use the next higher standard value if this not a standard value. in this example 511 ? is a standard value. check: because we must use a standard value for rcs and r slc; may be set at a different level (if the calculated value isn?t a standard value) and we must calculate the actual value (remember is the same as ). limit pkl i _ limit pkl i _ limit pkin _ limit pkl i _ i rearranging equation 14a to solve for : limit pkl i _ cs slc ramp pkin r dri i limit ) 45.0( _ ? = a ua i limit actualin 34.2 150. )75.0511 25045.0( _ = ? = this is higher than the initial limit because we have to use standard values for r cs a i pkl 9.1 2.1 max__ =
micrel, inc. MIC3230/1/2 january 2009 15 m9999-011409-a and for r slc . if is too high than use a higher value for r cs . the calculated value of r cs for a 1.9a current limit was 179m ? . in this example, we have chosen a lower value which results in a higher current limit. if we use a higher standard value the current limit will have a lower value. the designer does not have the same choices for small valued resistors as with larger valued resistors. the choices differ from resistor manufacturers. if too large a current sense resistor is selected, the maximum output power may not be able to be achieved at low input line voltage levels. make sure the inductor will not saturate at the actual current limit . limit actualin i _ a 51178.0 :uu limit actualin i _ v pinis 250 _ perform a check at i in =2.34apk. v ma 45.015034.2 :u p maximum power dissipated in r cs is; eq. (17) cs r rms u 2 rr ip cs cs _ eq. (18) ? ? 1 ? _ max_ rms fet i rms 12 2 _ 2 max__ max_ _ ppl avein i id cs r i rmsa rms _44.1 12 26.0 64.178.0 2 2 ? ? 1 ? i cs r _ watt p cs r 31.015.25.1 2 u use a 1/2 watt resistor for r cs . output capacitor in this led driver application, the iled ripple current is a more important factor compared to that of the output ripple voltage (although the two are directly related). to find the c out for a required iled ripple use the following calculation: for an output ripple 20% of ripple iled nom iled ma iled ripple 7035.02.0 u eq. (19) ) (* ** _ totalled adj ripple nom nom out rr iled td iled c find the equivalent ac resistance from the datasheet of the led. this is the inverse slope of the iled vs. v f curve i.e.: acled r _ eq. (20) iled v r f acled ' ' _ in this example use for each led. : 1.0 _ acled r if the leds are connected in series, multiply s, we obtain the following: : 1.0 _ acled r by the total number of leds. in this example of 6 led : :u 6.01.06 _ totalled r uf rr iled td iled c totalled adj ripple nom nom out 1.4 ) (* ** _ use the next highest standard value, which is 4.7uf. e is shown in figure 5. for superior there is a trade off between the output ripple and th rising edge of the pwmd pulse. this is because between pwm dimming pulses, the converter stops pulsing and c out will start to dischar ge. the amount that c out will discharge depends on the time between pwm dimming pluses. at the next pwmd pulse c out has to be charged up to the full output voltage v out before the desired led current flows. input capacitor the input current performance, ceramic capacitors should be used because of their low equivalent series resistance (esr). the input ripple current is equal to the ripple in the inductor plus the ripple voltage across the input capacitor, which is the esr of c in times the inductor ripple. the input capacito r will also bypass the emi generated by the converter as well as any voltage spikes generated by the inductance of the input line. for a required v in_ripple : eq. (21) f khz mv a f v i c sw ripple in ppin in p 4.1 500 508 28.0 8 _ _ uu u u this is the minimum value that should be used. the e, the fet has to hold off an output input capacitor should also be rated for the maximum rms input current. to protect the ic from inductive spikes or any overshoot, a larger value of input capacitance may be required and it is recommended that ceramic capacitors be used. in this design example a value of 4.7f ceramic capacitor was selected. mosfet selection in this design exampl voltage maximum of 30v. it is recommended to use an 80% de-rating value on switching fets, so a minimum of a 38v fet should be selected. in this design example, a 75v fet has been selected. the switching fet power losses are the sum of the conduction loss and the switching loss: eq. (22) fet cond fet fet p pp _ _ switch the conduct ion loss of the fet is when the fet is , where turned on. the conduction power loss of the fet is found by the following equation: eq. (23) fet cond fet i p _ _ dson rms r u 2
micrel, inc. MIC3230/1/2 january 2009 16 m9999-011409-a ? ? ? ? ? ? ? ? + = 12 2 _2 _ _ ppl avein rmsfet i id i the switching losses occur during the switching transitions of the fet. the transition times, t transition , are ere are the times when the fet is turning off and on. th two transition times per period, t. it is important not to confuse t (the period) with the transition time, t transition . eq. (24) fsw t 1 = eq. (25) sw transition out ave fet switch fet f t v i p = max_ max_ max__ max_ _ max_ transition t : to find eq. (26) igatedrv qg t transition max_ is the total gate c rge of the external mosfet provided by the mosfet manufacturer and charge at a where qg ha the qg ould chosen at a v gs 10v. this is not an exact value, but is more of an estimate of max_ transition t . the fet manufacturers? provide a gate specified v gs voltage: sh gs g in q c _ = fet v @ this is the fet?s input capacitance. select a fet with r ds(on) and q g such that the external power is below harge=68nc (typical) the perature. as the erature at 125c is: ation 23: about 0.7w for a so-8 or about 1w for a powerpak (fet package). the vishay siliconix si7148dp in a powerpak so-8 package is one good choice. the internal gate driver in the MIC3230/1/2 is 2a. from the si7148dp data sheet: r ds(on)_25c =0.0145 ? total gate c ) is a function of tem ( ( temp r onds ) in the fet increases so does the temp r ds(on) . to find )( )( temp r onds use equation 27, or simply o o double the ) for )125( )( c r onds . eq. (27) ( onds r 25( )( c r onds ) 007.1()25( )( )25( ) )( o o ? = temp onds c rtemp the ( r ds )( ) temp on ? ? ? m c r dson 30)7 )125( )25125( o o = 00.1(0145.0 from equ mw m p cond fet 62 3064.1 2 _ =?= from equation 26: ns a nc igatedrv qg t transition 34 2 68 == a ave 64.1 max__ = i fet v v out 28 max_ = from equation 25: watts khz nsva 78.0 500342864.1 p switch fet max_ _ = = from equation 22 w w mw 84.078.062 =+ p fet = this about the limit for a part on a circuit board without having to use any additional heat sinks. is best used here because of the lower d the low reverse recovery time. the rectifier diode a schottky diode forward voltage an voltage stress on the diode is the max v out and therefore a diode with a higher rating than max v out should be used. an 80% de-rating is recommended here as well. eq (28) ? ? ? ? ? 2 _ diode i i ? ? ? + ?= 12 )1( _ 2 max__ max_ ppl avein rms id eq. (29) w p ivp diode rms diode schottky diode 81.0 max__ m ic3230 power losses ic3230are: is the total gate charge of the the power losses in the m eq.(30) fvq p gate gate mic = 3230 vini q + where gat q external e t. r ati mosfe gate v is the gate drive voltage of the MIC3230. f is the switching frequency. q i is the quiescent cu rent of the MIC3230 found in th lectrical characteriz on table. mai q 2.3 = . v in is the voltage at the v in pin of the MIC3230 . from eq.(30) ma khz nf p mic 142.35001268 3230 e e w 45.0 = + =
micrel, inc. MIC3230/1/2 january 2009 17 m9999-011409-a ovp - over voltage protection 2. even though the rrc is very short (tens of nanoseconds) the peak currents are high (multiple amperes). the high rrc causes a voltage drop on the ground trace of the pcb and if the converter control ic is referenced to this voltage drop, the output regulation will suffer. set ovp higher than the maximum output voltage by at least one volt. to find the resistor divider values for ovp use equation 3 and set the ovp=30v and r8=100k ? : : u: k k r 33.4 245.130 245.1100 9 3. it is important to connec t the ic?s reference to the same point as the output capacitors to avoid the voltage drop caused by rrc. this is also called a star connection or single point grounding. pcb layout 1. all typologies of dc-to- dc converters have a reverse recovery current (rrc) of the flyback or (freewheeling) diode. even a schottky diode, which is advertised as having zero rrc, it really is not zero. the rrc of the freewheeling diode in a boost converter is even greater than in the buck converter. this is because the output voltage is higher than the input voltage and the diode has to charge up to ?v out during each on-time pulse and then discharge to v f during the off-time. 4. feedback trace: the high impedance traces of the fb should be short.
micrel, inc. MIC3230/1/2 january 2009 18 m9999-011409-a package information 10-pin msop (mm)
micrel, inc. MIC3230/1/2 january 2009 19 m9999-011409-a 12-pin 3mm 3mm mlf ? (ml)
micrel, inc. MIC3230/1/2 january 2009 20 m9999-011409-a 16-pin exposed pad tssop (tse) micrel, inc. 2180 fortune drive san jose, ca 95131 usa tel +1 (408) 944-0800 fax +1 (408) 474-1000 web http://www.micrel.com the information furnished by micrel in this data sheet is believed to be accurate and reliable. however, no responsibility is a ssumed by micrel for its use. micrel reserves the right to c hange circuitry and specificati ons at any time without notification to the customer. micrel products are not designed or authorized for use as components in life support appliances, devices or systems where malfu nction of a product reasonably be expected to result in pers onal injury. life support devices or system s are devices or systems that (a) are in tended for surgical impla into the body or (b) support or sustain life, and whose failure to perform can be reasonably expect ed to result in a significan t injury to the user. a purchaser?s use or sale of micrel produc ts for use in life support app liances, devices or systems is a purchaser?s own risk and purchaser agrees to fully indemnify micrel for any damages resu lting from such use or sale. can nt ? 2009 micrel, incorporated.

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